Method and apparatus for filtering signals utilizing a vibrating micromechanical resonator

ABSTRACT

Several MEMS-based methods and architectures which utilize vibrating micromechanical resonators in circuits to implement filtering, mixing, frequency reference and amplifying functions are provided. Apparatus is provided for filtering signals utilizing vibrating micromechanical resonators. One of the primary benefits of the use of such architectures is a savings in power consumption by trading power for high selectivity (i.e., high Q). Consequently, the present invention relies on the use of a large number of micromechanical links in SSI networks to implement signal processing functions with basically zero DC power consumption.

CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This application is a continuation-in-part of copending U.S.patent application entitled “Device Including A MicromechanicalResonator Having An Operating Frequency And Method Of Extending Same”filed Jan. 13, 2000 and having U.S. Ser. No. 09/482,670 which, in turn,claims the benefit of U.S. provisional application entitled “VHFFree-Free Beam High-Q Micromechanical Resonators”, filed Jan. 14, 1999and having U.S. Ser. No. 60/115,882. This application also claims thebenefit of U.S. provisional application entitled “Transceiver Front-EndArchitectures Using Vibrating Micromechanical Signal Processors” filedApr. 20, 2000 and having U.S. Ser. No. 60/199,063.

[0002] This invention was made with government support under ContractNo. F30602-97-2-0101 awarded by DARPA. The government has certain rightsin the invention.

BACKGROUND OF THE INVENTION

[0003] 1. Field of the Invention

[0004] This invention relates to methods and apparatus for filteringsignals utilizing a vibrating micromechanical resonator.

[0005] 2. Background Art

[0006] The need for passive off-chip components has long been a keybarrier against communication transceiver miniaturization. Inparticular, the majority of the high-Q bandpass filters commonly used inthe RF and IF stages of heterodyning transceivers are realized usingoff-chip, mechanically-resonant components, such as crystal and ceramicfilters and SAW devices, as illustrated in FIG. 1. Due to higher qualityfactor Q, such technologies greatly outperform comparable filtersimplemented using transistor technologies, in insertion loss, percentbandwidth, and achievable rejection. High Q is further required toimplement local oscillators or synchronizing clocks in transceivers,both of which must satisfy strict phase noise specifications. Again, asillustrated in FIG. 1, off-chip elements (e.g., quartz crystals) areutilized for this purpose.

[0007] Being off-chip components, the above mechanical devices mustinterface with integrated electronics at the board level, and thisconstitutes an important bottleneck against the miniaturization ofsuper-heterodyne transceivers. For this reason, recent attempts toachieve single-chip transceivers for paging and cellular communicationshave utilized alternative architectures that attempt to eliminate theneed for off-chip high-Q components via higher levels of transistorintegration. Unfortunately, without adequate front-end selectivity, suchapproaches have suffered somewhat in overall performance, to the pointwhere they so far are usable only in less demanding applications.

[0008] Given this, and recognizing that future communication needs willmost likely require higher levels of performance, single-chiptransceiver solutions that retain high-Q components and that preservesuper-heterodyne-like architectures are desirable.

[0009] Recent demonstrations of vibrating beam micromechanical(“μmechanical”) resonator devices with frequencies in the VHF range andQ's in the tens of thousands have sparked a resurgence of researchinterest in communication architectures using high-Q passive devices asdisclosed in the above-noted patent application entitled “DeviceIncluding A Micromechanical Resonator Having An Operating Frequency andMethod of Extending Same.” Much of the interest in these devices derivesfrom their use of IC-compatible microelectromechanical systems (MEMS)fabrication technologies to greatly facilitate the on-chip integrationof ultra-high-Q passive tanks together with active transistorelectronics, allowing substantial size reduction.

[0010]FIG. 2 illustrates a comparison of MEMS and SAW technologieswherein MEMS offers the same or better high-Q frequency selectivity withorders of magnitude smaller size. Indeed, reductions in size andboard-level packaging complexity, as well as the desire for the highperformance attainable by super-heterodyne architectures, are principaldrivers for this technology.

[0011] Although size reduction is certainly an advantage of thistechnology (commonly dubbed “RF MEMS”), it merely touches upon a muchgreater potential to influence general methods for signal processing. Inparticular, since they can now be integrated (perhaps on a massivescale) using MEMS technology, vibrating μmechanical resonators (orμmechanical links) can now be thought of as tiny circuit elements, muchlike resistors or transistors, in a new mechanical circuit technology.Like a single transistor, a single mechanical link does not possessadequate processing power for most applications. However, again liketransistors, when combined into larger (potentially, VLSI) circuits, thetrue power of μmechanical links can be unleashed, and signal processingfunctions with attributes previously inaccessible to transistor circuitsmay become feasible.

[0012] The Need for High Q in Oscillators

[0013] For any communications application, the stability of theoscillator signals used for frequency translation, synchronization, orsampling, is of utmost importance. Oscillator frequencies must be stableagainst variations in temperature against aging, and against anyphenomena, such as noise or microphonics, that cause instantaneousfluctuations in phase and frequency. The single most important parameterthat dictates oscillator stability is the Q of the frequency-settingtank (or of the effective tank for the case of ring oscillators). For agiven application, and assuming a finite power budget, adequate long-and short-term stability of the oscillation frequency is insured onlywhen the tank Q exceeds a certain threshold value.

[0014] Given the need for low power in portable units, and given thatthe synthesizer (containing the reference and VCO oscillators) is oftena dominant contributor to total transceiver power consumption, moderntransceivers could benefit greatly from technologies that yield high-Qtank components.

[0015] The Need for High O in Filters

[0016] Tank Q also greatly influences the ability to implement extremelyselective IF and RF filters with small percent bandwidth, small shapefactor, and low insertion loss. As tank Q decreases, insertion lossincreases very quickly, too much even for IF filters, and quiteunacceptable for RF filters. As with oscillators, high-Q tanks arerequired for RF and IF filters alike, although more so for the latter,since channel selection is done predominantly at the IF insuper-heterodyne receivers. In general, the more selective the filter,the higher the resonator Q required to achieve a given level ofinsertion loss.

[0017] Micromechanical Circuits

[0018] Although mechanical circuits, such as quartz crystal resonatorsand SAW filters, provide essential functions in the majority oftransceiver designs, their numbers are generally suppressed due to theirlarge size and finite cost. Unfortunately, when minimizing the use ofhigh-Q components, designers often trade power for selectivity (i.e.,Q), and hence, sacrifice transceiver performance. As a simpleillustration, if the high-Q IF filter in the receive path of acommunication subsystem is removed, the dynamic range requirement on thesubsequent IF amplifier, IQ mixer, and A/D converter circuits, increasesdramatically, forcing a corresponding increase in power consumption.Similar trade-offs exist at RF, where the larger the number or greaterthe complexity of high-Q components used, the smaller the powerconsumption in surrounding transistor circuits.

[0019] The Micromechanical Beam Element

[0020] To date, the majority of μmechanical circuits most useful forcommunication applications in the VHF range have been realized usingμmechanical flexural-mode beam elements, such as shown in FIG. 2 withclamped-clamped boundary conditions. Although several micromachiningtechnologies are available to realize such an element in a variety ofdifferent materials, surface micromachining has been the preferredmethod for μmechanical communication circuits, mainly due to itsflexibility in providing a variety of beam end conditions and electrodelocations, and its ability to realize very complex geometries withmultiple levels of suspension.

[0021] U.S. Pat. No. 6,049,702 to Tham et al. discloses an integratedpassive transceiver section wherein microelectromechanical (MEM) devicefabrication techniques are used to provide low loss, high performanceswitches. Utilizing the MEM devices also makes possible the fabricationand use of several circuits comprising passive components, therebyenhancing the performance characteristics of the transceiver.

[0022] U.S. Pat. No. 5,872,489 to Chang et al. discloses an integratedtunable inductance network and method. The network utilizes a pluralityof MEM switches which selectively interconnect inductance devicesthereby providing a selective inductance for a particular circuit.

[0023] U.S. Pat. No. 5,963,857 to Greywall discloses an articlecomprising a micromachined filter. In use, the micromachined filters areassembled as part of a radio to miniaturize the size of the radio.

[0024] U.S. Pat. Nos. 5,976,994 and 6,169,321 to Nguyen et al. disclosea batch-compatible, post-fabrication annealing method and system to trimthe resonance frequency and enhance the quality factor ofmicromechanical structures.

[0025] U.S. Pat. Nos. 5,455,547; 5,589,082 and 5,537,083 to Lin et al.disclose microelectromechanical signal processors. The signal processorsinclude many individual microelectromechanical resonators which enablethe processor to function as a multi-channel signal processor or aspectrum analyzer.

[0026] U.S. Pat. No. 5,640,133 to MacDonald et al. discloses acapacitance-based, tunable, micromechanical resonator. The resonatorsmay be selectively tuned and used in mechanical oscillators,accelerometers, electromechanical filters and other electronic devices.

[0027] U.S. Pat. Nos. 5,578,976 to Yao, 5,619,061 to Goldsmith et al.and 6,016,092 to Qiu et al. disclose various micromechanical andmicroelectromechanical switches used in communication apparatus.

[0028] U.S. Pat. No. 5,839,062 to Nguyen et al. disclose a MEMS-basedreceiver including parallel banks of microelectromechanical filters.

[0029] U.S. Pat. Nos. 5,491,604 and 5,955,932 to Nguyen et al. discloseQ-controlled microresonators and tunable filters using the resonators.

[0030] U.S. Pat. No. 5,783,973 to Weinberg et al. discloses amicromechanical, thermally insensitive silicon resonator and oscillator.

[0031] The following articles are of general interest: Nguyen et al.,“Design and Performance of CMOS Micromechanical Resonator Oscillators”,1994 IEEE INTERNATIONAL FREQUENCY CONTROL SYMPOSIUM, Pp. 127-134; Wanget al, “Q-Enhancement of Microelectromechanical Filters Via Low-VelocitySpring Coupling”, 1997 IEEE ULTRASONICS SYMPOSIUM, pp. 323-327; Bannon,III et al., “High Frequency Microelectromechanical IF Filters”, 1996IEEE ELECTRON DEVICES MEETING, San Francisco, Calif., Dec.8-11, 1996,pp. 773-776; and Clark et al., “Parallel-Resonator HF MicromechanicalBandpass Filters” 1997 INTERNATIONAL CONFERENCE ON SOLID-STATE SENSORSAND ACTUATORS”, pp. 1161-1164.

[0032] U.S. Pat. No. 5,640,133 to MacDonald et al. discloses acapacitance-based tunable micromechanical resonator. The resonatorincludes a movable beam which holds a plurality of electrodes. Theresonator also includes a plurality of stationary electrodes. Inoperation, an adjustable bias voltage, applied to the beam electrodesand the stationary electrodes, is used to adjust the resonant frequencyof the resonator.

[0033] U.S. Pat. No. 5,550,516 to Burns et al. discloses an integratedresonant microbeam sensor and transistor oscillator. The sensor andoscillator, capable of providing high-Q values, utilizes variouscircuitry, electrode placement, and various configurations of microbeamgeometry to vary the operating resonant frequency.

[0034] U.S. Pat. No. 5,399,232 to Albrecht et al. discloses amicrofabricated cantilever stylus with an integrated pyramidal tip. Thepyramidal tip, integrally formed on the cantilever arm, limits themovement of the arm in the direction of the tip.

[0035] U.S. Pat. No. 4,262,269 to Griffin et al. discloses a Q-enhancedresonator which utilizes resonator positioning to provide a desiredperformance. Resonators are separated by one-quarter-wavelengthdistances to obtain desired loss characteristics.

[0036] U.S. Pat. No. 4,721,925 to Farace et al. discloses amicromechanical electronic oscillator etched from a silicon wafer. Thepatent discusses the configuration and the circuitry which enables theoscillator to perform according to desired characteristics.

[0037] The following U.S. patents are generally related to thisinvention: U.S. Pat. Nos. 4,081,769; 4,596,969; 4,660,004; 4,862,122;5,065,119; 5,191,304, 5,446,729; 5,428,325, 5,025,346; 5,090,254;5,455,547; 5,491,604; 5,537,083; and 5,589,082.

SUMMARY OF THE INVENTION

[0038] An object of the present invention is to provide a method andapparatus for filtering signals utilizing a vibrating micromechanicalresonator wherein the resonator is isolated from a support structure forthe resonator during resonator vibration.

[0039] In carrying out the above object and other objects of the presentinvention, a method is provided for filtering signals to obtain adesired passband of frequencies. The method includes providing amicromechanical filter apparatus including a micromechanical resonatorhaving a fundamental resonant mode formed on a substrate and a supportstructure anchored to the substrate to support the resonator above thesubstrate. The method also includes vibrating the resonator so that theapparatus passes a desired frequency range of signals whilesubstantially attenuating signals outside the desired frequency range.The support structure is attached to the resonator so that the resonatoris isolated from the support structure during resonator vibration.

[0040] The step of vibrating may include forcing different portions ofthe resonator to move in opposite directions at the same time so thatthe resonator vibrates in a resonant mode, m, higher than thefundamental resonant mode wherein the resonator has m+1 nodal points.

[0041] The micromechanical filter apparatus may include a plurality ofinput electrodes spaced along the resonator to allow electrostaticexcitation of the resonator. The step of forcing may include the stepsof applying an in-phase signal to one of the input electrodes to deflecta first portion of the resonator in a first direction and applying anout-of-phase signal to another input electrode to deflect a secondportion of the resonator in a second direction opposite the firstdirection to force the resonator into a correct mode shape.

[0042] The micromechanical filter apparatus may include an inputelectrode formed on the substrate to allow electrostatic excitation ofthe resonator. The step of forcing may include the step of applying asignal to the input electrode. The resonator and the input electrode maydefine a capacitive transducer gap therebetween. The resonator mayfurther include m+1 spacers having a height and which extend between theresonator and the substrate at the m+1 nodal points. The m+1 spacersforce the resonator into a correct mode shape during the application ofthe signal to the input electrode.

[0043] Further, in carrying out the above object and other objects ofthe present invention a micromechanical filter apparatus for filteringsignals to obtain a desired passband of frequencies is provided. Theapparatus includes a substrate, a plurality of intercoupledmicromechanical elements including a resonator and a support structureanchored to the substrate to support the elements above the substrate.The support structure and the resonator are both dimensioned so that theresonator is isolated from the support structure during resonatorvibration. Energy losses to the substrate are substantially eliminated.The apparatus is a high-Q apparatus.

[0044] The support structure may be attached to the resonator at atleast one nodal point of the resonator.

[0045] The signals may be RF signals.

[0046] The apparatus may be an RF filter apparatus.

[0047] The apparatus may be a bandpass filter apparatus.

[0048] The support structure may include at least one beam attached to anodal point of the resonator.

[0049] The apparatus may further include at least one input electrodeformed on the substrate to allow electrostatic excitation of theresonator wherein the resonator and the at least one input electrodedefine a capacitive transducer gap therebetween.

[0050] The apparatus may further include at least one spacer having aheight. Each spacer extends between the resonator and the substrate at anodal point of the resonator. The size of the gap is based on the heightof the at least one spacer during pull down of the resonator.

[0051] The apparatus may be a silicon-based filter apparatus.

[0052] The apparatus may be a diamond-based filter apparatus.

[0053] The apparatus may further include at least one output electrodeformed on the substrate to sense output of the apparatus.

[0054] The support structure may include a plurality of beams and theresonator may include a plurality of nodal points. Each of the beams isattached to the resonator at one of the nodal points of the resonator sothat the resonator sees substantially no resistance to transverse ortorsional motion from the support structure.

[0055] A pair of balanced input electrodes may be formed on thesubstrate to allow electrostatic excitation of the resonator.

[0056] A pair of balanced output electrodes may be formed on thesubstrate to sense output of the apparatus.

[0057] The plurality of intercoupled micromechanical elements mayinclude a pair of intercoupled end resonators.

[0058] The support structure may support the end resonators above thesubstrate.

[0059] The plurality of intercoupled micromechanical elements mayinclude an inner resonator intercoupled to the end resonators.

[0060] The support structure may support the end and inner resonatorsabove the substrate.

[0061] The plurality of intercoupled micromechanical elements mayfurther include a plurality of coupling links for coupling the innerresonator to the end resonators.

[0062] The coupling links may be operable in multiple modes.

[0063] The coupling links may be higher mode coupling beams.

[0064] The above object and other objects, features, and advantages ofthe present invention are readily apparent from the following detaileddescription of the best mode for carrying out the invention when takenin connection with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0065]FIG. 1 is a prior art, schematic view of the front-end of atransceiver including off-chip, board-level implementation of SAW,ceramic and crystal resonators in a schematic perspective view;

[0066]FIG. 2a is a prior art, schematic perspective view of a SAWresonator and a number of MEMS resonators formed on a silicon die tocompare the two approaches;

[0067]FIG. 2b is a prior art, enlarged schematic perspective view of oneof the MEMS resonators as indicated at 2 b in FIG. 2a;

[0068]FIG. 3 is a system level schematic block diagram of the front-enddesign for a typical wireless transceiver showing off-chip, high-Q,passive components targeted for replacement via micromechanical versionsof the present invention;

[0069]FIG. 4 is a graph of transmission [dB] versus frequencyillustrating desired filter characteristics;

[0070]FIG. 5a is a perspective schematic view of a symmetricaltwo-resonator VHF μmechanical filter with typical bias, excitation andsignal conditioning electronics;

[0071]FIG. 5b is an electrical equivalent circuit for the filter of FIG.5a;

[0072]FIG. 6 is a system level block diagram of an RF front-end receiverincluding an RF channel-select receiver architecture utilizing largenumbers of micromechanical resonators in banks and schematically andperspectively illustrating a typical micromechanical filter of FIG. 5a;

[0073]FIG. 7 is a system/circuit diagram for an RF channel-selectmicromechanical filter bank;

[0074]FIG. 8 is a system level block diagram of the RF front-endreceiver of FIG. 6 and schematically and perspectively illustrating amicromechanical switch thereof;

[0075]FIG. 9 is a system level block diagram of the RF front-endreceiver of FIGS. 6 and 8 and schematically and perspectivelyillustrating a mixer-filter-gain stage thereof based on the filter ofFIGS. 6 and 5a;

[0076]FIG. 10 is a system level block diagram of the RF front-endreceiver of FIGS. 6, 8 and 9 and schematically and perspectivelyillustrating a micromechanical resonator oscillator thereof;

[0077]FIG. 11 is a system/circuit diagram for a switchable μmechanicalresonator synthesizer;

[0078]FIG. 12 is a system level block diagram of the RF front-endreceiver of FIGS. 6, 8, 9 and 10 wherein the LNA is shown eliminated byphantom lines due to the low loss channel selector, the T/R switch andthe mixer-filter-gain stage;

[0079]FIG. 13 is a system block diagram architecture showing the receivepath of a communication device;

[0080]FIG. 14 is a system block diagram of an RF channel-selecttransmitter architecture utilizing high-power μmechanical resonators;

[0081]FIG. 15 is a schematic top perspective view of a 92 MHz (VHF)free-free beam polysilicon μmechanical resonator wherein support beamsisolate the resonator beam element from a substrate thereby allowinghigher Q operation;

[0082]FIG. 16 is a graph illustrating a measured frequencycharacteristic for the resonator of FIG. 15;

[0083]FIG. 17a is a schematic perspective view of a UHF μmechanicalfilter utilizing free-free beam μmechanical resonators designed tooperate at a second mode; and

[0084]FIG. 17b is a partial equivalent circuit for the filter of FIG.17a, identifying the circuit functions of individual beam elements.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT(S)

[0085] To illustrate more concretely the specific transceiver functionsthat can benefit from micromechanical implementations to be discussedherein, FIG. 3 presents a system level schematic block diagram for atypical super-heterodyne wireless transceiver. A small box is positionedin the corner of each box to represent a component that can be replacedwith a micromechanical (MEMS) version. As implied in FIG. 3, several ofthe constituent components can already be miniaturized using integratedcircuit transistor technologies. These include the low noise amplifiers(LNA's) in the receive path, the solid-state power amplifier (SSPA) inthe transmit path, synthesizer phase-locked loop (PLL) electronics,mixers, and lower frequency digital circuits for baseband signaldemodulation. Due to noise, power, and frequency considerations, theSSPA (and sometimes the LNA's) are often implemented using compoundsemiconductor technologies (i.e., GaAs). Thus, they often occupy theirown chips, separate from the other mentioned transistor-basedcomponents, which are normally realized using silicon-based bipolar andCMOS technologies. However, given the rate of improvement of silicontechnologies (silicon-germanium included), all of the above functionsmay be integrated onto a single-chip.

[0086] Unfortunately, placing all of the above functions onto a singlechip does very little toward decreasing the overall super-heterodynetransceiver size, which is dominated not by transistor-based components,but by the numerous passive components indicated in FIGS. 1 and 3. Thepresence of so many frequency-selective passive components is easilyjustified when considering that communication systems designed toservice large numbers of users require numerous communication channels,which in many implementations (e.g., Time Division Multiple Access(TDMA)) must have small bandwidths and must be separable by transceiverdevices used by the system. The requirement for small channel bandwidthsresults in a requirement for extremely selective filtering devices forchannel selection and extremely stable (noise-free) local oscillatorsfor frequency translation. For the vast majority of cellular andcordless standards, the required selectivity and stability can only beachieved using high-Q components, such as discrete inductors, discretetunable capacitors (i.e., varactors), and SAW and quartz crystalresonators, all of which interface with IC components at the boardlevel. The needed performance cannot be achieved using conventional ICtechnologies, because such technologies lack the required Q. It is forthis reason that virtually all commercially available cellular orcordless phones contain numerous passive SAW and crystal components.

[0087] Micromechanical Resonators

[0088] For communications applications, clamped-clamped and free-freeflexural-mode beams with Q's on the order of 10,000 (in vacuum) andtemperature coefficients on the order of −12 ppm/° C., are available forthe VHF range, while thin-film bulk acoustic resonators (Q˜1,000) haveso far addressed the UHF range.

[0089] From a design perspective, one Q-limiting loss mechanism thatbecomes more important with increasing frequency is loss to thesubstrate through anchors. The frequency dependence of this mechanismarises because the stiffness of a given resonator beam generallyincreases with resonance frequency, giving rise to larger forces exertedby the beam on its anchors during vibration. As a consequence, moreenergy per cycle is radiated into the substrate via the anchors.Anti-symmetric resonance designs, such as balanced tuning forks, couldprove effective in alleviating this source of energy loss.

[0090] Anchor loss mechanisms can be greatly alleviated by using“anchor-less” resonator designs, such as shown in the above-noted patentapplication Ser. No. 09/482,670 and as illustrated by the view of FIG.15. This device utilizes a free-free beam (i.e., xylophone) 60 suspendedabove a ground plane and sense electrode 61 and a drive electrode 63 byfour torsional support beams 62 attached at flexural node points. Thebeams 62, in turn, are supported at anchors 64. By choosing supportdimensions corresponding to a quarter-wavelength of the free-free beam'sresonance frequency, the impedance presented to the beam 60 by thesupports 62 can be effectively nulled out, leaving the beam 60 virtuallylevitated and free to vibrate as if it had no supports. UHF frequency isobtained via use of free-free beam resonators specifically designed tooperate at higher modes. The desired mode is selected (while suppressingother modes), by strategic placement and excitation of electrodes, andby the use of dimples under the structure 60 to force node locationscorresponding to the desired mode. The mode can also be specified byplacing the support beams at nodal locations, as in FIG. 17a.

[0091]FIG. 16 shows a frequency characteristic for a 92.25 MHz versionof this μmechanical resonator with a Q of nearly 8,000—still plenty forchannel-select RF applications.

[0092] Table 1 presents expected resonance frequencies for various beamdimensions, modes, and structural materials, showing a wide range ofattainable frequencies, from VHF to UHF. TABLE 1 μMechanical ResonatorFrequency Design* Frequency (MHz) Material Mode h_(r) [μm] W_(r) [μm]L_(r) [μm] 70 silicon 1 2 8 14.54 110 silicon 1 2 8 11.26 250 silicon 12 4 6.74 870 silicon 2 2 4 4.38 1800 silicon 3 1 4 3.09 1800 diamond 3 14 6.16

[0093] Micromechanical Filters

[0094] Among the more useful μmechanical circuits for communications arethose implementing low-loss bandpass filters, capable of achievingfrequency characteristics as shown in FIG. 4 where a broader frequencypassband than achievable by a single resonator beam is shown, with asharper roll-off to the stopband (i.e., smaller shape factor).

[0095] To achieve the characteristic of FIG. 4, a number ofmicromechanical resonators may be coupled together by soft couplingsprings. By linking resonators together using (ideally) masslesssprings, a coupled resonator system is achieved that exhibits severalmodes of vibration. The frequency of each vibration mode corresponds toa distinct peak in the force-to-displacement frequency characteristic,and to a distinct, physical mode shape of the coupled mechanicalresonator system. For the example case of a three-resonator filter, inthe lowest frequency mode, all resonators vibrate in phase; in themiddle frequency mode, the center resonator ideally remains motionless,while the end resonators vibrate 180° out of phase; and finally, in thehighest frequency mode, each resonator is phase-shifted 180° from itsadjacent neighbor. Without additional electronics, the completemechanical filter exhibits a jagged passband. As described hereinbelow,termination resistors designed to lower the Q's of the input and outputresonators by specific amounts are required to flatten the passband andachieve a more recognizable filter characteristic, such as in FIG. 4.

[0096] The filters use a number of high-Q micromechanical beam elementsconnected in a network that achieves the specified bandpass frequencyresponse. If effect, a micromechanical filter is another example of amicromechanical circuit, similar to that of FIG. 15, but in this caseusing a plurality of beam elements to achieve a frequency shapingresponse not achievable by a single beam element.

[0097] In practical implementations, because planar IC processestypically exhibit substantially better matching tolerances thanabsolute, the constituent resonators in μmechanical filters are normallydesigned to be identical, with identical dimensions and resonancefrequencies. For such designs, the center frequency of the overallfilter is equal to the resonance frequency ƒ_(o) of the resonators,while the filter passband (i.e., the bandwidth) is determined by thespacings between the mode peaks.

[0098] The relative placement of the vibration peaks in the frequencycharacteristic—and thus, the passband of the eventual filter—isdetermined primarily by the stiffnesses of the coupling springs(k_(sij)) and of the constituent resonators at the coupling locations(k_(r)). Specifically, for a filter with center frequency ƒ_(o) andbandwidth B, these stiffnesses must satisfy the expression:$\begin{matrix}{B = {\left( \frac{f_{o}}{k_{ij}} \right)\left( \frac{k_{sij}}{k_{r}} \right)}} & (1)\end{matrix}$

[0099] where k_(ij) is a normalized coupling coefficient found in filtercookbooks. The filter bandwidth is not dependent on the absolute valuesof resonator and coupling beam stiffness; rather, their ratiok_(sij)/k_(r) dictates bandwidth. Thus, the procedure for designing amechanical filter involves two main steps (not necessarily in thisorder): first, design of a mechanical resonator with resonance frequencyƒ_(o) and adjustable stiffness k_(r); and second, design of couplingsprings with appropriate values of stiffness k_(sij) to enable a desiredbandwidth within the adjustment range of resonator k_(r)'s.

[0100] To take advantage of the maturity of LC ladder filter synthesistechniques, the enormous database governing LC ladder filterimplementations, and the wide availability of electrical circuitsimulators, realization of a particular μmechanical filter often alsoinvolves the design of an LC ladder version to fit the desiredspecification. The elements in the LC ladder design are then matched tolumped mechanical equivalents via electromechanical analogy, whereinductance, capacitance, and resistance in the electrical domain equateto mass, compliance, and damping, respectively, in the mechanicaldomain.

[0101] A Two-Resonator HF-VHF Micromechanical Filter

[0102]FIG. 5a shows a perspective view schematic of a practicaltwo-resonator micromechanical filter capable of operation in the HF toVHF range. As shown, the filter consists of two μmechanicalclamped-clamped beam resonators with anchors 18 at their opposite ends,coupled mechanically by a soft coupling spring or beam 19, all suspendedabove a substrate (not shown). Conductive (polysilicon) strips 20, 22,24, and 26 underlie each resonator by approximately 1000 Å (as also inFIGS. 6, 9 and 17 a), a center one 20 serving as a capacitive transducerinput electrode positioned to induce resonator vibration in a directionperpendicular to the substrate, a center one 24 serving as an outputelectrode and the flanking ones 22 and 26 serving as tuning or frequencypulling electrodes capable of voltage-controlled tuning of resonatorfrequencies. The resonator-to-electrode gaps are determined by thethickness of a sacrificial oxide spacer during fabrication and can thusbe made quite small (e.g., 0.1 μm or less) to maximize electromechanicalcoupling.

[0103] The filter is excited with a DC-bias voltage V_(P) applied to theconductive mechanical network, and an AC signal applied to the inputelectrode, but this time through an appropriately valued sourceresistance R_(Q) that loads the Q of the input resonator to flatten thepassband. The output resonator of the filter must also see a matchedimpedance to avoid passband distortion, and the output voltage ν_(o) isgenerally taken across this impedance. As described hereinbelow, therequired value of I/O port termination resistance can be tailored fordifferent applications, and this can be advantageous when designing lownoise transistor circuits succeeding the filter, since such circuits canthen be driven by optimum values of source resistance to minimize noise.

[0104] From a signal flow perspective, the operation of the above filtercan be briefly summarized as follows:

[0105] (1) An electrical input signal is applied to the input port andconverted to an input force by the electromechanical transducer (whichfor the case of FIG. 5a is capacitive) that can then induce mechanicalvibration in the x direction;

[0106] (2) Mechanical vibration comprises a mechanical signal that isprocessed in the mechanical domain—specifically, the signal is rejectedif outside the passband of the filter, and passed if within thepassband; and

[0107] (3) The mechanically processed signal appears as motion of theoutput resonator and is reconverted to electrical energy at the outputtransducer, ready for processing by subsequent transceiver stages.

[0108] From the above, the name “micromechanical signal processor”clearly suits this device. Details of the design procedure formicromechanical filters now follow.

[0109] HF-VHF Filter Design

[0110] As can be summarized from FIG. 5b, the network topologies for themechanical filters of this work differ very little from those of theirpurely electronic counterparts, and in principal, can be designed at thesystem-level via a procedure derived from well-known, coupled resonatorladder filter synthesis techniques. In particular, given the equivalentLCR element values for a prototype μmechanical resonator, it is possibleto synthesize a mechanical filter entirely in the electrical domain,converting to the mechanical domain only as the last step. However,although possible, such a procedure is not recommended, since knowledgeand ease of design in both electrical and mechanical domains can greatlyreduce the effort required.

[0111] The design procedure for the two-resonator micromechanical filterof FIG. 5a can be itemized as follows:

[0112] (1) Design and establish the μmechanical resonator prototype tobe used, choosing necessary geometries for the needed frequency andinsuring that enough electrode-to-resonator transducer coupling isprovided to allow for predetermined termination resistor values. For agiven resonator, with predetermined values of W_(r), h, W_(e), V_(P),and R_(Q), this amounts to solving for the resonator length L_(r) andelectrode-to-resonator gap spacing d that simultaneously satisfy anumber of well known equations. Table 2 summarizes the needed gapspacings to achieve various values of R_(Q) for micromechanical filterscentered at 70 MHz and 870 MHz, and with Q=10,000, B=1.25 MHz, andV_(P)=10V. TABLE 2 Two-Resonator μMechanical FilterElectrode-to-Resonator Gap Spacing Design* Gap Spacing, d, for R_(Q) =Frequency 300Ω 500Ω 1,000Ω 2,000Ω 5,000Ω  70 MHz^(†) 195Å 223Å 266Å 317Å399Å 870 MHz^(‡) 78Å 81Å 80Å 95Å 119Å

[0113] (2) Choose a manufacturable value of coupling beamwidth W_(s12)and design coupling beam(s) corresponding to a “quarter-wavelength” ofthe filter center frequency. Here, the coupling beam is recognized as anacoustic transmission line that can be made transparent to the filterwhen designed with quarter-wavelength dimensions as described in theprior art.

[0114] (3) Determine the coupling location(s) on the resonatorscorresponding to the filter bandwidth of interest. This procedure isbased upon two important properties of this filter and the resonatorscomprising it: First, the filter bandwidth B is determined not byabsolute values of stiffness, but rather by a ratio of stiffnesses(k_(s12)/k_(rc)), where the subscript c denotes the value at thecoupling location; and second, the value of resonator stiffness k_(rc)varies with location (in particular, with location velocity) and so canbe set to a desired value by simply choosing an appropriate couplingbeam attachment point. The location is easily determined as described inthe prior art.

[0115] (4) Generate a complete equivalent circuit for the overall filterand verify the design using a circuit simulator. FIG. 5b presents theequivalent circuit for the two-resonator micromechanical filter of FIG.5a. Each of the outside resonators are modeled via circuits. Thecoupling beam actually operates as an acoustic transmission line, andthus, is modeled by a T-network of energy storage elements.

[0116] Transformers are used between the resonator and coupling beamcircuits of FIG. 5b to model the velocity transformations that arisewhen attaching the coupling beams at locations offset from the center ofthe resonator beam. The whole circuit structure of FIG. 5b can berecognized as that of the LC ladder network for a bandpass filter.

[0117] Further details on the design of micromechanical filters can befound in the literature.

[0118] RF Micromechanical Filters

[0119] As shown in FIG. 16, one of the highest demonstrated frequenciesto date for polysilicon micromechanical resonators is 92 MHz with aQ˜8,000. The above-noted patent application Ser. No. 09/482,670 alsodiscloses a way of extending the frequency. As shown in Table 1, theabove frequency (and higher) is geometrically feasible, but specialdesign and material precautions are necessary to maintain adequate Q asfrequencies rise. If Q's can be maintained >5,000, then record insertionloss performance on the order of 0.5 dB should be achievable, whichcould greatly enhance the sensitivity of receivers used in both short-and long-range communications. In fact, if Q˜10,000 can be achieved,then RF channel-selection would be achievable using a switchable bank offilters, one for each channel, as described hereinbelow. The ability todo channel-selection right at the front-end of a transceiver canpotentially save substantial amounts of power in both the receive andtransmit paths, due to relaxed dynamic range requirements and theability to use higher efficiency power amplifiers.

[0120]FIG. 17a is similar to FIG. 5a and presents the schematic of afilter structure, along with a partial equivalent electrical circuit ofFIG. 17b (obtained via electromechanical analogy) that identifies themechanical network as a bandpass filter. In particular, the filterstructure is seen to be comprised of a number of mechanical resonators(modeled by LCR tanks) connected by acoustic transmission lines (modeledby T-networks of energy storage elements)—a structure similar to otherresonator-based filters, but using micromechanical elements with ordersof magnitude higher Q, giving it the ability to perform with much lowerinsertion loss than other technologies. (Not to mention orders ofmagnitude smaller size.)

[0121] In the design of FIG. 17a, UHF frequency is obtained via use of asecond mode free-free beam resonator including beams 70 and 72 coupledtogether and to an output beam 74 by coupling beams 73. The resonator isspecifically designed to operate at higher modes. The desired mode isselected (while suppressing other modes), by strategic placement andexcitation of balanced input electrodes 76 and 78, and by the use ofspacers or dimples 80 under the beams 70, 72 and 74 at flexural nodalpoints to force node locations corresponding to the desired mode.

[0122] Clearly for each mode, m, there exist m+1 nodal points offlexural deflection. One technique for exciting these higher modes isthe use of differential signaling with the input electrodes 76 and 78.An in-phase signal may be applied to the electrode 76 that induces beamdeflection in one direction, while an out-of-phase signal is applied tothe electrode 78 that induces beam deflection in the opposite directionas dictated by the mode shape. This technique for second mode excitationis extended readily to any mode of vibration.

[0123] As an additional yield- and Q-enhancing feature, the capacitorgap spacing in this device is not entirely determined via a thinsacrificial oxide, as was done in previous clamped-clamped beam highfrequency devices. Rather, a capacitor gap is now determined by theheight of the dimples 80, set via a timed etch. The height of thedimples 80 is such that when a sufficiently large DC-bias is appliedbetween the input electrodes 76 and 78 and the resonator beam 70 andbetween the output electrodes 81 and the beam 74, the whole structurecomes down and rests upon the dimples 80, which are located at theflexural nodal points, and thus, have little impact on filter operation.The dimples 80 may be formed either on the beams 70, 72 and 74 or on thesubstrate.

[0124] The input electrodes 76 and 78 allow electrostatic excitation viaan applied AC voltage. Balanced output electrodes 81 are positionedunder the output beam 74. The output electrodes 81 senses or detectoutput currents off the beam 74.

[0125] The beams 70 and 74 are supported by non-intrusive supports orbeams 82 and 84, respectively, which, in turn, are supported by anchors86 and 88, respectively. While not illustrated, one support beam can beused for each beam 70 and 74 to further minimize energy dissipation.Alternatively one support beam can support all of the beams 70, 72 and74.

[0126] The beams 82 and 84 are strategically designed with dimensions,so as to effect an impedance transformation that isolates the beams 70and 74 from the rigid anchors 86 and 88, respectively. Ideally, thebeams 70 and 74 see zero-impedance into their supports or beams 82 and84, respectively, and thus, effectively operate as if levitated withoutany supports. As a result, anchor dissipation mechanisms normally foundin previous clamped-clamped beam resonators are greatly suppressed,allowing much higher apparatus Q.

[0127] The filter of FIG. 17a constitutes the first attempt of its kindto implement filter circuits using free-free beam micromechanicalresonators. Unlike previous filters using clamped-clamped beams (thatcould only attain VHF frequency), both transversal and torsional motionsare considered for the coupling beams in FIG. 17a.

[0128] Because the beams 82 and 84 are attached at the node points, thesupport springs or beams 82 and 84 (ideally) sustain no translationalmovement during resonator vibration, and thus, support (i.e., anchor)losses due to translational movements—such as those sustained byclamped-clamped beam resonators—are greatly alleviated. Furthermore,with the recognition that the supporting torsional beams 82 and 84actually behave like acoustic transmission lines at the frequencies ofinterest, torsional loss mechanisms can also be negated by strategicallychoosing support dimensions so that they present virtually no impedanceto the beams 70 and 74. As a result, the beams 70 and 74 effectively“see” no supports at all and operate as if levitated above thesubstrate, devoid of anchors and their associated loss mechanisms.

[0129] Micromechanical Mixer-Filters

[0130]FIG. 9 presents the schematic for a symmetrical μmechanicalmixer-filter, showing the bias and input scheme required fordown-conversion with a gain stage. As shown, since this device providesfiltering as part of its function, the overall mechanical structure isexactly that of a μmechanical filter. The only differences are theapplied inputs and the use of a non-conductive coupling beam to isolatethe IF port from the LO. If the source providing V_(P) to the secondresonator is ideal (with zero source resistance) and the seriesresistance in the second resonator is small, LO signals feeding acrossthe coupling beam capacitance are shunted to AC ground before reachingthe IF port. In reality, finite resistivity in the resonator materialallows some amount of LO-to-IF leakage.

[0131] The mixer conversion gain/loss in this device is determinedprimarily by the relative magnitudes of the DC-bias V_(P) applied to theresonator and the local oscillator amplitude V_(LO).

[0132] In general, conversion gain is possible if V_(LO)>V_(P).

[0133] Micromechanical Switches

[0134] The mixer-filter device described above is one example of amechanical circuit that harnesses non-linear device properties toprovide a useful function. Another very useful mode of operation thatfurther utilizes the non-linear nature of the device is a μmechanicalswitch. FIG. 8 presents an operational schematic for a μmechanicalswitch. A conductive or actuation plate 30 is suspended above a pair ofactuation electrodes 32 by suspension beams 34 having anchors 36. Aswitch conductor portion 38 of the plate 30 is suspended over a pair ofgrounds 40 and a sense electrode or conductor 42. When the switch is inthe “on-state” here, the conductor 42 is shorted to the grounds 40. Theoperation of the switch of FIG. 8 is fairly simple: To achieve the“on-state” for one of the electrodes apply a sufficiently large voltageacross the plate and a desired electrode to pull that part of the platedown and short it (in either a DC or AC fashion) to the desiredelectrode.

[0135] In general, to minimize insertion loss, the majority of switchesuse metals as their structural materials. It is their metal constructionthat makes μmechanical switches so attractive, allowing them to achieve“on-state” insertion losses down to 0.1 dB—much lower than FETtransistor counterparts, which normally exhibit ˜2 dB of insertion loss.In addition to exhibiting such low insertion loss, μmechanical switchesare extremely linear, with IIP3's greater than 66 dBm, and can bedesigned to consume no DC power (as opposed to FET switches, which sinka finite current when activated).

[0136] RF Receiver Front-End Architectures Using MEMS

[0137] The methods by which the above-noted mechanical circuits are bestincorporated into communications sub-systems are now considered. Threeapproaches to using micromechanical vibrating resonators are describedin order of increasing performance enhancement:

[0138] 1) Direct replacement of off-chip high-Q passives;

[0139] 2) Use of an RF channel select architecture using a large numberof high-Q micromechanical resonators in filter banks and switchablenetworks; and

[0140] 3) Use of an all-mechanical RF front-end.

[0141] In the RF channel-select architecture, μmechanical circuits areassumed to be able to operate at UHF with Q's on the order of 10,000.However, this isn't absolutely necessary. It is needed for today'scommunications, but in the future, a communications standard may comeabout that allows even less Q.

[0142] Direct Replacement of Off-Chip High-Q Passives

[0143] Perhaps the most direct way to harness μmechanical circuits isvia direct replacement of the off-chip ceramic, SAW, and crystalresonators used in RF preselect and image reject filters, IFchannel-select filters, and crystal oscillator references, asillustrated in FIGS. 1 and 3. In addition to high-Q components, FIG. 3also shows the use of other MEMS-based passive components, such asmedium-Q micromachined inductors and tunable capacitors used in VCO'sand matching networks, as well as low-loss (˜0.1 dB) μmechanicalswitches that not only provide enhanced antenna diversity, but that canalso yield power savings by making TDD (rather than FDD) more practicalin future transceivers.

[0144] Of course, the main benefits from the above approach to usingMEMS are size reduction and, given the potential for integration of MEMSwith transistor circuits, the ability to move more components onto thesilicon die. A limited number of performance benefits also result fromreplacement of existing high-Q passives by μmechanical ones, such as theability to tailor the termination impedances required by RF and IFfilters (c.f., Table 2). Such impedance flexibility can be beneficialwhen designing low-noise amplifiers (LNA's) and mixers in CMOStechnology, which presently often consume additional power to impedancematch their outputs to 50Ω off-chip components. If higher impedances canbe used, for example at the output of an LNA, significant power savingsare possible. As an additional benefit, since the source impedancepresented to the LNA input is now equal to R_(Q), it can now be tailoredto minimize noise figure (NF).

[0145] Although beneficial, the performance gains afforded by meredirect replacement by MEMS are quite limited when compared to moreaggressive uses of MEMS technology. More aggressive architectures aredescribed hereinbelow.

[0146] An RF Channel-Select Architecture

[0147] To fully harness the advantages of μmechanical circuits, one mustfirst recognize that due to their micro-scale size and zero DC powerconsumption, μmechanical circuits offer the same system complexityadvantages over off-chip discrete components that planar IC circuitsoffer over discrete transistor circuits. Thus, to maximize performancegains, μmechanical circuits should be utilized on a massive scale, or atleast as much as possible.

[0148] Perhaps one of the simplest ways to harness the small size ofmicromechanical circuits is to add multi-band reconfigurability to atransceiver by adding a preselect and image reject filter for eachcommunication standard included. Due to the small size ofmicromechanical filters, this can be done with little regard to theoverall size of the transceiver.

[0149] Although the above already greatly enhances the capability oftoday's wireless transceivers, it in fact only touches upon a muchgreater potential for performance enhancement. In particular, it doesnot utilize micromechanical circuits to their fullest complexity. FIG. 6presents the system-level block diagram for a possible receiverfront-end architecture that takes full advantage of the complexityachievable via μmechanical circuits, such as the micromechanical filterof FIG. 5a. The main driving force behind this architecture is powerreduction, attained in several of the blocks by trading power for highselectivity (i.e., high-Q). The key power saving blocks in FIG. 6 arenow described.

[0150] Switchable RF Channel-Select Filter Bank

[0151] If channel selection (rather than pre-selection) were possible atRF frequencies (rather than just an IF), then succeeding electronicblocks in the receive path (e.g., LNA, mixer) would no longer need tohandle the power of alternate channel interferers. Thus, their dynamicrange can be greatly relaxed, allowing substantial power reductions. Inaddition, the rejection of adjacent channel interferers also allowsreductions in the phase noise requirements of local oscillator (LO)synthesizers, providing further power savings.

[0152] To date, RF channel selection has been difficult to realize viapresent-day technologies. In particular, low-loss channel selection atRF would require tunable resonators with Q's in the thousands.Unfortunately, however, high-Q often precludes tunability, making RFchannel selection via a single RF filter a very difficult prospect.

[0153] On the other hand, it is still possible to select individual RFchannels via many non-tunable, or slightly tunable, high-Q filters, onefor each channel, and each switchable (and tunable) by command.Depending upon the standard, this could entail hundreds or thousands offilters—numbers that would be absurd if off-chip macroscopic filters areused, but that may be perfectly reasonable for microscale, passive,μmechanical filters, such as previously described.

[0154]FIG. 7 presents one fairly simple rendition of the key systemblock that realizes the desired RF channel selection. As shown, thisblock consists of a bank of μmechanical filters with all filter inputsconnected to a common block input and all outputs to a common blockoutput, and where each filter passband corresponds to a single channelin the standard of interest. However, it is to be noted that it does nothave to be limited to a single channel. It could also be severalchannels as well (e.g., 3 channels) and this could still be veryadvantageous, depending upon the communications standard.

[0155] In the scheme of FIG. 7, a given filter is switched on (with allothers off) by decoder-controlled application of an appropriate DC-biasvoltage to the desired filter. The desired force input and outputcurrent are generated in a μmechanical resonator only when a DC-biasV_(P) is applied (i.e., without V_(P), the input and output electrodesare effectively open-circuited).

[0156] The potential benefits afforded by this RF channel selector canbe quantified by assessing its impact on the LNA linearity specificationimposed by the IS-98-A interim standard for CDMA cellular mobilestations. In this standard, the required IIP3 of the LNA is set mainlyto avoid desensitization in the presence of a single tone (generated byAMPS) spaced 900 kHz away from the CDMA signal center frequency. Here,reciprocal mixing of the local oscillator phase noise with the 900 kHzoffset single tone and cross-modulation of the single tone with leakedtransmitter power outputs dictate that the LNA IIP3 exceeds +7.6 dBm.However, if an RF channel-select filter bank such as shown in FIG. 7precedes the LNA and is able to reject the single tone by 40 dB, therequirement on the LNA then relaxes to IIP3 ≦−29.3 dBm (assuming thephase noise specification of the local oscillator is not also relaxed).Given the well known noise and linearity versus power trade-offsavailable in LNA design, such a relaxation in IIP3 can result in nearlyan order of magnitude reduction in power. In addition, since RF channelselection relaxes the overall receiver linearity requirements, it maybecome possible to put more gain in the LNA to suppress noise figure(NF) contributions from later stages, while relaxing the required NF ofthe LNA itself, leading to further power savings.

[0157] Turning to oscillator power, if the single tone interferer isattenuated to 40 dB, then reciprocal mixing with the local oscillator isalso greatly attenuated, allowing substantial reduction in the phasenoise requirement of the local oscillator. Requirement reductions caneasily be such that on-chip solutions to realization of the receive pathVCO (e.g., using spiral inductors and pn-diode tunable capacitors)become plausible.

[0158] Practical implementations of the switchable filter bank requiremultiplexing support electronics that must interconnect with eachμmechanical device. If implemented using a two-chip approach, the numberof chip-to-chip bonds required could become quite cumbersome, making asingle-chip solution desirable.

[0159] In the pursuit of single-chip systems, several technologies thatmerge micromachining processes with those for integrated circuits havebeen developed and implemented over the past several years. Thesetechnologies can be categorized into three major approaches: mixedcircuit and micromechanics, pre-circuits, and post-circuits. Thesetechnologies, however, are well known in the art and are not discussedherein.

[0160] Switchable Micromechanical Resonator Synthesizer

[0161] Although the μmechanical RF channel selector described above withreference to FIGS. 5, 6 and 7 may make possible the use of existingon-chip technologies to realize the receive path VCO as shown in FIG. 3,this approach is not recommended, since it denies the system fromachieving much greater power reduction factors that may soon beavailable through MEMS technology. In particular, given that power and Qcan often be interchanged when designing for a given oscillator phasenoise specification, a better approach to implementing the VCO would beto use μmechanical resonators (with orders of magnitude higher Q thanany other on-chip tank) to set the VCO frequency. In fact, with Q's ashigh as achievable via μmechanics, the basic design methodologies foroscillators must be re-evaluated. For example, in the case where theoscillator and its output buffer contribute phase noise according toLeeson's equation, where the 1/ƒ²-to-white phase noise corner occurs at(ƒ_(o)(2Q)), a tank Q>1,500 is all that would be required to move the1/ƒ²-to-white phase noise corner close enough to the carrier that onlywhite phase noise need be considered for CDMA cellular applications,where the phase noise power at frequency offsets from 285 kHz to 1515kHz is most important. If only white noise is important, then only theoutput buffer noise need be minimized, and sustaining amplifier noisemay not even be an issue. If so, the power requirement in the sustainingamplifier might be dictated solely by loop gain needs (rather than byphase noise needs), which for a μmechanical resonator-based VCO withR_(x) ˜40Ω, L_(x) 84 μH, and C_(x) ˜0.5ƒf, might be less than 1 mW.

[0162] To implement a tunable local oscillator synthesizer, a switchablebank, similar to that of FIG. 7 but using μmechanical resonators 46, notfilters, each corresponding to one of the needed LO frequencies, andeach switchable into or out of an oscillator sustaining circuit 48 bytransistor switches 49 at their electrodes 51 is illustrated in FIG. 11and is preferred over the tuning-fork-resonator oscillator 50illustrated in FIG. 10. Because μmechanical resonators are now used inthis implementation, the Q and thermal stability of the oscillator maynow be sufficient to operate without the need for locking to a lowerfrequency crystal reference. The power savings attained upon removingthe PLL and prescaler electronics needed in past synthesizers canobviously be quite substantial. In effect, by implementing thesynthesizer using μmechanical resonators, synthesizer power consumptioncan be reduced from the ˜90 mW dissipated by present-day implementationsusing medium-Q L and C components, to something in the range of only 1-4mW. Again, all this is attained using a circuit topology that would seemabsurd if only macroscopic high-Q resonators were available, but thatbecomes plausible in the micromechanical arena.

[0163]FIG. 11 presents the basic topology of the LO synthesizer. Here,rather than tune a single medium Q resonator such as done in present-dayVCO's, a bank of numerous switchable high-Q (Q>5,000) micromechanicalresonators is utilized, where each resonator corresponds to one of theneeded frequencies in a given communications network. In this scheme, nophase-locking circuit is required, since the local oscillator merelyswitches the appropriate resonator into the oscillator feedback loop togenerate the needed output frequency. Because the Q of themicromechanical resonators are orders of magnitude higher than before(Q˜5,000, as opposed to 40),the phase noise performance of thisoscillator should be orders of magnitude better than present-day VCO's.In fact, with such a high Q, the 1/ƒ²-to-white noise corner frequency ofthe standard phase noise vs. frequency plot may be so close to thecarrier, that 1/ƒ² noise may no longer be a major consideration, andonly white noise is present at the important frequency offsets. In thiscase, power could be traded for Q to such an extent that it may becomepossible to operate the oscillator with less than 1 mW of powerconsumption. Combined with the fact that phase-locking is no longerrequired in the scheme of FIG. 11, this constitutes more than 90 mW ofpower savings, while improving the performance of the oscillator.

[0164] It should be emphasized that the oscillator scheme of FIG. 11would be difficult to practically realize without MEMS technology—inparticular, without its ability to attain Q's>5,000 in such a tiny size.Specifically, present-day LO synthesizers are constrained to use low-Qtanks in their VCO's because such tanks are much more tunable than theirhigh-Q counterparts; i.e., high-Q generally implies low tunability.Thus, to attain tunability using high-Q resonators, many resonators arerequired, one for each frequency to be generated. This, of course, wouldrequire an absurdly large volume if macroscopic high-Q components areutilized. With μmechanical high-Q elements, however, hundreds or eventhousands of resonators can fit within a 0.5×0.5 cm² area, so thesynthesizer architecture of FIG. 11 becomes plausible.

[0165] A single-chip LO synthesizer, using the architecture of FIG. 11can achieve impressive phase noise performance at UHF frequencies (˜800MHz). By using higher-mode, free-free beam resonator designs previouslydescribed, UHF (and possibly S-Band or higher) frequency synthesizersare feasible. K- or Ka-Band applications are not unreasonable, andshould benefit from the architecture of FIG. 11 either directly asshown, or indirectly, since a UHF high-Q reference oscillator shouldgreatly improve the stability of K- or Ka-band synthesizers.

[0166] Micromechanical Mixer-Filter

[0167] The use of a μmechanical mixer-filter in the receive path asillustrated in FIG. 9 eliminates the DC power consumption associatedwith the active mixer normally used in present-day receivearchitectures. The mixer-filter-gain stage includes a pair of n-typeresonators 21,23 coupled together by a p-type or undoped beam or spring19 similar to filters of FIGS. 5a and 6.

[0168] This corresponds to a power savings on the order of 10-20- mW. Inaddition, if multiple input electrodes (one for RF, one for matching)are used for the mixer-filter, the RF input can be made to appear purelycapacitive to the LNA (i.e., at the RF frequency), and the LNA would nolonger require a driver stage to match a certain impedance. Rather, aninductive load can be used to resonate the capacitance, allowing powersavings similar to that previously discussed in association withimpedance matching.

[0169] An All-MEMS RF Front-End Receiver Architecture

[0170] In the above MEMS-based architecture, if μmechanical filters andmixer-filters can post insertion losses consistent with their high-Qcharacteristics, then an LNA is not really required at RF frequencies,as illustrated by phantom lines for the LNA in FIG. 12. FIG. 13 depictsa receive path comprised of a relatively wideband image rejectμmechanical RF filter followed immediately by a narrowband IFmixer-filter that then feeds subsequent IF electronics. The only activeelectronics operating at RF in this system are those associated with thelocal oscillator, which if it uses a bank of μmechanical resonators aspreviously described, may be able to operate at less than 1 mW. Thearchitecture of FIG. 13 clearly presents enormous power advantages,eliminating completely the power consumption of the LNA and active mixerof FIG. 3—a total power savings on the order of 40 mW—and together withthe 90 mW of power savings from the micromechanical LO, substantiallyincreasing mobile phone standby times.

[0171] An RF Transmitter Architecture Using MEMS

[0172] Due to a lack of sufficient in-band power handling capability,very little consideration has been given to date to the possibility ofusing μmechanical resonators in the transmit path.

[0173]FIG. 14 depicts one rendition, in which an RF channel selector isplaced after the power amplifier (PA) in the transmit path. This channelselector uses a similar circuit as that of FIG. 7, but using μmechanicalresonators with sufficient power handling capability. This transmittopology provides enormous power savings. In particular, the high-Q,high-power filter with less than 1 dB of insertion loss follows the PA,cleaning all spurious outputs, including those arising from spectralregrowth. Consequently, more efficient PA designs can be utilized,despite their non-linearity. For example, a PA previously restricted bylinearity considerations to 30% efficiency in present-day transmitterarchitectures, may now be operable closer to its maximum efficiency,perhaps 50%. For a typical transmit power of 600 mW, this efficiencyincrease corresponds to 800 mW of power savings. If a more efficient PAtopology could be used, such as Class E, with theoretical efficienciesapproaching 100%, the power savings could be much larger.

[0174] In addition to the MEMS-based channel-select RF filter bank, thearchitecture of FIG. 14 also features a micromechanical upconverter thatuses a mixer-filter device, such as previously described, to upconvertand filter the information signal before directing it to the poweramplifier.

[0175] There are two preferred methods for using the structure of theupper part of FIG. 9 as an upconverting device. In the first method, thebaseband signal at frequency ω_(IF) to be upconverted is applied to theinput electrode 20 and the upconverting carrier signal at frequencyω_(LO) is applied to the input resonator device 21. Upconversion occursthrough mixing around the nonlinear capacitive transducer between theinput electrode 20 and the resonator 21. Specifically, the electricalbaseband information signal at frequency ω_(IF) is upconverted to aforce at frequency ω_(LO)+ω_(IF). This force is then filtered by thefilter structure of FIG. 9, which is now designed to have a passbandaround ω_(LO)+ω_(IF). If this passband is made small enough, channelselection to the point of removing not only distortion harmonics, butalso spectrally regrown components, is possible. By placing a dc-biasV_(P) on the output resonator 23, the displacement of this resonator isconverted to an electrical output voltage or current, depending upon theoutput load. It should be noted, also, that gain is possible inupconverting the baseband signal to the RF signal, so this stage alsoserves as a gain stage, as well.

[0176] The second method for upconversion, involves filtering the basedband signal first, then upconverting. In this method, the basebandsignal is again applied to the input electrode 20, but the dc-bias isapplied to the input resonator, and the carrier signal to the outputresonator. This way, the baseband signal is first filtered by thestructure, then upconverted at the output via electromechanicalamplitude modulation. Again, gain is possible in this configuration.

[0177] High power handling micromechanical resonators may usealternative geometries (e.g., no longer flexural mode) and the use ofalternative transduction (e.g., piezoelectric, magnetostrictive).

[0178] Summary

[0179] Vibrating μmechanical resonators constitute the building blocksfor a new integrated mechanical circuit technology in which high Qserves as a principal design parameter that enables more complexcircuits. By combining the strengths of integrated μmechanical andtransistor circuits, using both in massive quantities, previouslyunachievable functions become possible that enable transceiverarchitectures with projections for orders of magnitude performancegains. In particular, with the addition of high-Q μmechanical circuits,paradigm-shifting transceiver architectures that trade power forselectivity (i.e., Q) become possible, with the potential forsubstantial power savings and multi-band reconfigurability.

[0180] While the best mode for carrying out the invention has beendescribed in detail, those familiar with the art to which this inventionrelates will recognize various alternative designs and embodiments forpracticing the invention as defined by the following claims.

What is claimed is:
 1. A method for filtering signals to obtain adesired passband of frequencies, the method comprising: providing amicromechanical filter apparatus including a micromechanical resonatorhaving a fundamental resonant mode formed on a substrate and a supportstructure anchored to the substrate to support the resonator above thesubstrate; and vibrating the resonator so that the apparatus passes adesired frequency range of signals while substantially attenuatingsignals outside the desired frequency range, wherein the supportstructure is attached to the resonator so that the resonator is isolatedfrom the support structure during resonator vibration.
 2. The method asclaimed in claim 1 wherein the step of vibrating includes forcingdifferent portions of the resonator to move in opposite directions atthe same time so that the resonator vibrates in a resonant mode, m,higher than the fundamental resonant mode wherein the resonator has m+1nodal points.
 3. The method as claimed in claim 2 wherein themicromechanical filter apparatus includes a plurality of inputelectrodes spaced along the resonator to allow electrostatic excitationof the resonator and wherein the step of forcing includes the steps ofapplying an in-phase signal to one of the input electrodes to deflect afirst portion of the resonator in a first direction and applying anout-of-phase signal to another input electrode to deflect a secondportion of the resonator in a second direction opposite the firstdirection to force the resonator into a correct mode shape.
 4. Themethod as claimed in claim 2 wherein the micromechanical filterapparatus includes an input electrode formed on the substrate to allowelectrostatic excitation of the resonator and wherein the step offorcing includes the step of applying a signal to the input electrode,the resonator and the input electrode defines a capacitive transducergap therebetween and wherein the micromechanical resonator furtherincludes m+1 spacers having a height and which extend between theresonator and the substrate at the m+1 nodal points and wherein the m+1spacers force the resonator into a correct mode shape during theapplication of the signal to the input electrode.
 5. A micromechanicalfilter apparatus for filtering signals to obtain a desired passband offrequencies, the apparatus comprising: a substrate; a plurality ofintercoupled micromechanical elements including a resonator; and asupport structure anchored to the substrate to support the elementsabove the substrate wherein the support structure and the resonator areboth dimensioned so that the resonator is isolated from the supportstructure during resonator vibration wherein energy losses to thesubstrate are substantially eliminated and wherein the apparatus is ahigh-Q apparatus.
 6. The apparatus as claimed in claim 5 wherein thesupport structure is attached to the resonator at at least one nodalpoint of the resonator.
 7. The apparatus as claimed in claim 5 whereinthe signals are RF signals.
 8. The apparatus as claimed in claim 7wherein the apparatus is an RF filter apparatus.
 9. The apparatus asclaimed in claim 5 wherein the apparatus is a bandpass filter apparatus.10. The apparatus as claimed in claim 5 wherein the support structureincludes at least one beam attached to a nodal point of the resonator.11. The apparatus as claimed in claim 5 further comprising at least oneinput electrode formed on the substrate to allow electrostaticexcitation of the resonator wherein the resonator and the at least oneinput electrode define a capacitive transducer gap therebetween.
 12. Theapparatus as claimed in claim 11 further comprising at least one spacerhaving a height, each spacer extending between the resonator and thesubstrate at a nodal point of the resonator wherein the size of the gapis based on the height of the at least one spacer during pull down ofthe resonator.
 13. The apparatus as claimed in claim 5 wherein theapparatus is a silicon-based filter apparatus.
 14. The apparatus asclaimed in claim 5 wherein the apparatus is a diamond-based filterapparatus.
 15. The apparatus as claimed in claim 11 further comprisingat least one output electrode formed on the substrate to sense output ofthe apparatus.
 16. The apparatus as claimed in claim 5 wherein thesupport structure includes a plurality of beams and the resonatorincludes a plurality of nodal points and wherein each of the beams isattached to the resonator at one of the nodal points of the resonator sothat the resonator sees substantially no resistance to transverse ortorsional motion from the support structure.
 17. The apparatus asclaimed in claim 11 wherein a pair of balanced input electrodes areformed on the substrate to allow electrostatic excitation of theresonator.
 18. The apparatus as claimed in claim 15 wherein a pair ofbalanced output electrodes are formed on the substrate to sense outputof the apparatus.
 19. The apparatus as claimed in claim 5 wherein theplurality of intercoupled micromechanical elements includes a pair ofintercoupled end resonators.
 20. The apparatus as claimed in claim 19wherein the support structure supports the end resonators above thesubstrate.
 21. The apparatus as claimed in claim 19 wherein theplurality of intercoupled micromechanical elements further includes aninner resonator intercoupled to the end resonators.
 22. The apparatus asclaimed in claim 21 wherein the support structure supports the end andinner resonators above the substrate.
 23. The apparatus as claimed inclaim 21 wherein the plurality of intercoupled micromechanical elementsfurther include a plurality of coupling links for coupling the innerresonator to the end resonators.
 24. The apparatus as claimed in claim23 wherein the coupling links are operable in multiple modes.
 25. Theapparatus as claimed in claim 23 wherein the coupling links are highermode coupling beams.